Wide dynamic range transimpedance amplifier with a controlled low frequency cutoff at high optical power

ABSTRACT

A wide dynamic range transimpedance amplifier with a low cut off frequency at high optical power. An automatic transimpedance gain control and DC cancellation control feedback circuit includes variable impedance circuitry. An emitter terminal of a first pnp transistor is connected to the input of the transimpedance amplifier. The impedance seen at the emitter terminal changes according to the average value of the input current. Open loop gain of the feedback loop including the first pnp transistor is not dependent on the average input current as the input current increases. A base terminal of the first pnp transistor is connected to a base terminal of second pnp transistor. Emitter size of the second pnp transistor is some factor N smaller than emitter size of the first pnp transistor. N can be configured to adjust the gain control and low corner frequency variation with input power.

RELATED APPLICATIONS

The present application is a continuation-in-part of U.S. patentapplication Ser. No. 10/371,847, filed Feb. 21, 2003, and entitled “AWide Range Transimpedance Amplifier With A Controlled Low FrequencyCutoff At High Optical Power, which is hereby incorporated by reference.That application claims the benefit of U.S. Provisional Application No.60/429,129, filed Nov. 26, 2002 and entitled “Circuit for Wide DynamicRange Transimpedance Amplifier,” which is hereby incorporated byreference.

BACKGROUND OF THE INVENTION

1. The Field of the Invention

The present invention relates to a wide dynamic range transimpedanceamplifier. More particularly, the present invention relates to a widedynamic range transimpedance amplifier with a controlled low cutofffrequency as optical power received at the transimpedance amplifierincreases.

2. The Relevant Technology

Fiber optic networks use light signals to transmit data over a network.Although light signals are used to carry data, the light signals aretypically converted into electrical signals in order to extract andprocess the data. The conversion of an optical signal into an electricalsignal is often achieved utilizing a fiber optic receiver. A fiber opticreceiver converts the optical signal received over the optical fiberinto an electrical signal, amplifies the electrical signal, and convertsthe electrical signal into an electrical digital data stream.

The fiber optic receiver usually includes a photodiode that detects thelight signal and converts the light signal into an electrical signal orcurrent. A transimpedance amplifier amplifies the signal from thephotodiode into a relatively large amplitude electrical signal. Theamplified electrical signal is then converted into a digital datastream.

The optical signals that are converted into electrical signals by thefiber optic receiver, however, can vary significantly in both amplitudeand power. The power of the optical signal is often related, forexample, to the length (hence loss) of the optical fiber over which theoptical signal was transmitted, the laser source power, the efficiencyof the photodiode, etc. These and other factors result in opticalsignals whose incident power at the transimpedance amplifier can varysignificantly.

Fiber optic receivers are only able to successfully receive and amplifyoptical signals that fall within a particular power range. In order fora fiber optic receiver to accommodate a wide range of optical signals,the fiber optic receiver and in particular, the transimpedanceamplifier, should be able to detect and amplify very low levels ofoptical power as well as high levels of optical power. The range ofsignals that can be successfully amplified is therefore effectivelylimited by the incident optical power because the fiber optic receiverdistorts or clamps signals whose optical power is too large and cannotrecognize signals whose optical power is too low.

One problem with current transimpedance amplifiers is that extending theability of the transimpedance amplifier to amplify signals with moreoptical power usually diminishes the ability of the transimpedanceamplifier to amplify signals with low optical power. In other words, themaximum optical input power that can be accepted by the transimpedanceamplifier while meeting signal integrity and bit error ratespecifications is usually specified as the input optical overload. Theminimum input power is specified as optical sensitivity. Thetransimpedance amplifier should be designed to maximize the opticaloverload and minimize the optical sensitivity. In most of the commercialor published transimpedance amplifiers, there is a direct tradeoffbetween the circuit optical (or current) sensitivity (or equivalentinput current noise) and the optical (or current) overload. Somesolutions to this problem, such as utilizing clamping circuitry orvoltage regulators to assist in the amplification of optical signalswith relatively large optical power, add both cost and complexity to thetransimpedance amplifier of the fiber optical receiver. Without the aidof additional circuitry, the range of signals that can be successfullyamplified is somewhat limited because the circuitry used for automaticgain control and DC cancellation introduces unwanted gain into thetransimpedance amplifiers DC cancellation feedback loop at large opticalpower.

The unwanted gain also has an adverse effect on the low frequency cutoffat higher optical powers. In other words, transimpedance amplifiers donot function at certain frequencies because the low frequency cutoff hasbeen increased. The low frequency cutoff for these types oftransimpedance amplifiers is related to the transconductance of thecircuitry used for automatic gain control and DC cancellation. Thus, asthe current of the input signal increases, the low frequency cutoff ofthe transimpedance amplifier is adversely affected.

BRIEF SUMMARY OF THE INVENTION

These and other limitations are overcome by the present invention, whichrelates to a wide range dynamic transimpedance amplifier. In the presentinvention, the wide dynamic range of the transimpedance amplifier isaccomplished in a manner where the gain in optical overload is notcompletely offset by a loss of optical sensitivity. In addition, the lowcutoff frequency does not increase linearly but approaches an upperlimit or is controlled as the input current to the transimpedanceamplifier increases. This permits, in one embodiment, the transimpedanceamplifier to be utilized with legacy systems that may operate at lowerfrequencies. The low cutoff frequency is controlled as the optical powerincreases.

In one embodiment, a transimpedance amplifier includes feedbackcircuitry that provides both automatic gain control, AC attenuation, DCshunting, and a low cutoff frequency at higher optical input powers. Apnp transistor is used in the DC cancellation feedback circuitry suchthat the emitter impedance of the pnp transistor is controlled, via afeedback loops by the average photodiode current. The emitter is alsoconnected with the input of the transimpedance amplifier.

As the photodiode current increases in response to increased opticalpower, the emitter impedance of the pnp transistor, which is connectedwith the input current or signal, decreases. However, unlike a commonemitter npn transistor whose collector is connected to the input of thetransimpedance amplifier, the pnp transistor does not introducesignificant additional gain into the feedback loop as the input signalamplitude increases, thereby keeping the low-cutoff frequencysubstantially unchanged.

An npn transistor can also be used, for example, when the transimpedanceamplifier is connected to the emitter of the npn transistor. Also, thenpn is used for in embodiments where a photodiode has its cathodeconnected to the input of the transimpedance amplifier.

Automatic gain control is achieved because the AC component of thephotodiode current is increasingly shunted to ground by the pnptransistor as the average photodiode current increases. The AC componentis attenuated at higher amplitudes. As the average photodiode currentdecreases, the emitter impedance of the pnp transistor decreases andenables low power signals to be passed with little or no attenuationinto the main amplifier. This ensures that the optical sensitivity ofthe transimpedance amplifier is not traded for optical overload. Inanother example, a shunt feedback transimpedance amplifier also includesfeedback circuitry to provide both automatic gain control, ACattenuation, and DC cancellation.

The variable impedance of the feedback circuitry can be achieved using apnp transistor, an npn transistor, field effect transistors, and thelike. In one embodiment, the emitter of an npn transistor is connectedwith an emitter of a pnp transistor such that current from thephotodiode can either be sourced or sunk. Photodiodes that amplify theinput current or signal can be accommodated by optimizing, in oneexample, the pnp transistor to trigger earlier.

In some embodiments of the invention, a transimpedance amplifierincludes a first pnp transistor and a second pnp transistor. The emitterterminal of the first pnp transistor is connected to the input of thetransimpedance amplifier. The base terminal of the first pnp transistoris connected to the base terminal of the second pnp transistor. Theemitter size of the second pnp transistor is some factor N smaller thanthe emitter size of the first pnp transistor.

Additional features and advantages of the invention will be set forth inthe description which follows, and in part will be obvious from thedescription, or may be learned by the practice of the invention. Thefeatures and advantages of the invention may be realized and obtained bymeans of the instruments and combinations particularly pointed out inthe appended claims. These and other features of the present inventionwill become more fully apparent from the following description andappended claims, or may be learned by the practice of the invention asset forth hereinafter.

BRIEF DESCRIPTION OF THE DRAWINGS

To further clarify the above and other advantages and features of thepresent invention, a more particular description of the invention willbe rendered by reference to specific embodiments thereof which areillustrated in the appended drawings. It is appreciated that thesedrawings depict only typical embodiments of the invention and aretherefore not to be considered limiting of its scope. The invention willbe described and explained with additional specificity and detailthrough the use of the accompanying drawings in which:

FIG. 1 illustrates an exemplary environment for implementing embodimentsof the present invention;

FIG. 2 is a block diagram of a transimpedance amplifier that providesboth automatic gain control and DC cancellation;

FIG. 3 illustrates one embodiment of the present invention in a commonbase configuration with a variable impedance formed using a pnptransistor;

FIG. 4 illustrates an embodiment of feedback circuitry in atransimpedance amplifier where the variable impedance includes both annpn transistor and a pnp transistor, thereby enabling the variableimpedance to either source or sink the current from the photodiode;

FIG. 5 depicts a shunt feedback transimpedance amplifier with automaticgain control and DC cancellation circuitry;

FIG. 6 illustrates another embodiment of feedback circuitry in atransimpedance amplifier using field effect transistors;

FIG. 7 plots the transimpedance of a transimpedance amplifier versus theaverage photodiode current;

FIG. 8 plots the low cutoff frequency of a transimpedance amplifierversus the average current of the photodiode;

FIG. 9 illustrates another embodiment of a transimpedance amplifier withautomatic gain control and DC cancellation circuitry;

FIG. 10 is an example plot of the gain of a transimpedance amplifierversus frequency;

FIG. 11 is an example of a number of plots of transimpedance amplifiertransimpedance gain versus photodiode average current; and

FIG. 12 is an example of a number of plots of transimpedance amplifierlow frequency cut-off versus photodiode average current.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT

The present invention relates to a wide dynamic range transimpedanceamplifier. The present invention more particularly relates to a widedynamic range transimpedance amplifier with automatic gain control anddirect current (DC) cancellation control. The automatic gain control anddirect current cancellation control are achieved in one embodiment usingvariable impedance circuitry whose impedance is controlled by or relatedto the average photodiode current. The variable impedance circuitry doesnot introduce significant open loop gain into the low frequency DC/AGCcancellation feedback loop of the transimpedance amplifier. In additionto automatic gain control and direct current cancellation, the opticalsensitivity of the transimpedance amplifier is not reduced while theoptical overload is increased.

As the average photodiode current increases, the impedance of thevariable impedance circuitry decreases. The variable impedance circuitrycancels the DC component of the input signal and attenuates the ACcomponent of the input signal, thereby providing automatic gain controlwhile canceling the DC component of the input signal.

FIG. 1 illustrates an exemplary environment for implementing embodimentsof the present invention. FIG. 1 illustrates a fiber optic receiver 100that receives an optical signal (light) and converts the optical signalto an electrical signal or data stream (usually represented as avoltage). The fiber optic receiver 100 receives an optical signal overan optical fiber 102. A photo diode 104 or other optical device thatconverts an optical signal to an electrical signal or current (light toelectrons conversion) receives the optical signal and generates anelectrical signal 110 (current). The transimpedance amplifier 120amplifies the electrical signal 110 to produce the amplified electricalsignal 112. The transimpedance amplifier 120 has a wide dynamic rangethat is able to amplify signals with large power without significantlydiminishing the ability to amplify signals with low power. The amplifiedelectrical signal 112 is then translated by the translation module 108and converted into an electrical digital signal 114.

FIG. 2 illustrates a block diagram of an exemplary transimpedanceamplifier in accordance with the present invention. The transimpedanceamplifier 120 includes an input stage 122 that receives an electricalcurrent 110 from a photo diode or other device that converts an opticalsignal into the electrical voltage. An amplifier 124 amplifies theelectrical signal and helps reduce or prevent noise from being a factor.A buffer 126 is also optionally provided at the output of thetransimpedance amplifier 120. In one embodiment, the input stage 122 andthe amplifier 124 are referred to as an forward transimpedance circuit.It is understood by one of skill in the art that the input stage 122 andthe amplifier 124 can be implemented in different configurations.Exemplary configurations include, but are not limited to, a common baseconfiguration and a shunt feedback configuration. In addition, theamplifier 124 includes single ended amplification, differentialamplification, and the like or any combination thereof. 341 The feedbackcircuit 130 provides both automatic gain control and direct current (DC)cancellation for the transimpedance amplifier 120. In the feedbackcircuit 130, a high frequency filter 132 is used to detect the DCcomponent output by the amplifier 124. The DC component or low frequencycomponent of the output of the amplifier 124 is passed by the highfrequency filter 132 and is canceled by the variable impedance circuitry140. In another embodiment, the high frequency filter 132 may bereplaced with a peak detector or similar circuitry.

The variable impedance circuitry 140 also provides automatic gaincontrol for the transimpedance amplifier 120 because it is able toattenuate at least some of the AC content of the photodiode current whenthe impedance of the variable impedance circuitry 140 decreases. Inother words, the impedance of the variable impedance circuitry 140changes according to the average current of the photodiode. As theaverage current received from the photodiode or other source increases,the impedance of the variable impedance circuitry decreases. Because theimpedance of the variable impedance circuitry 140 decreases, thevariable impedance circuitry 140 absorbs or attenuates some of the ACcomponent. This provides automatic control of the transimpedance gain ofthe fiber optic receiver. When the average photodiode current is low,the impedance of the variable impedance circuitry 140 is relativelylarge and the AC component is not absorbed or attenuated, but isamplified at the input stage 122 and/or by the amplifier 124. Thus, theoptical overload of the transimpedance amplifier is increased withoutsimultaneously trading off the optical sensitivity of the transimpedanceamplifier.

In This is advantageous for the transimpedance amplifier 120 because therange of signals that can be amplified without clipping, saturation, orother problems, is increased. Low power signals are also amplified bythe transimpedance amplifier 120 as the AC component is not absorbed orattenuated by the variable impedance circuitry 140, while opticalsignals with higher optical power are partially absorbed or attenuatedby the variable impedance circuitry 140. The transimpedance amplifier120 can thereby successfully amplify a wide range of signals.

FIG. 3 illustrates one embodiment of a transimpedance amplifier 120. Thetransimpedance amplifier 120 of FIG. 3 utilizes a common base topologywith a feedback circuit that provides both low frequency or DCcancellation and automatic gain control, as previously stated.Generally, the transimpedance amplifier includes an amplifier thatincludes one or more stages. The DC offset or component is sensed by thefeedback circuit and eliminated from the input signal. In the example ofFIG. 3, the transistors 200 and 202 are included in the input stage. Thecurrent from the photodiode is converted to a voltage by the transistor202. The voltage output by the transistor 200 serves as a referencevoltage in this embodiment. An output signal from the transistors 200and 202 is input to the transistors 206 and 208, which are arranged inan emitter follower configuration such that the voltage at the emitterssubstantially follows the voltage at the bases of the transistors 206and 208. The amplifier 210 amplifies the output of the emitter followers(206 and 208).

The DC or low frequency component of the output of the amplifier 210 ispassed by the low frequency operational amplifier 214, which is anexample of a high frequency filter, and drives the base of the pnptransistor 204. Also, the DC or low frequency component can be sensed atthe output of the input stage or at the output of the output of theemitter follower transistors 206 and 208.

FIG. 3, the transistor 204 is a pnp transistor and the DC component orlow frequency component detected by the low frequency operationalamplifier 214 drives the base of the pnp transistor 204. The emitter ofthe pnp transistor 204 is also electrically connected with the signalgenerated by the photodiode. As the average photodiode currentincreases, the emitter impedance of the transistor 204 decreases. Thisenables some of the AC component being processed by the transistor 202to be absorbed by the transistor 204 and permits the transimpedanceamplifier to amplify or transmit signals whose optical power is large.The transistor 204 is an example of the variable impedance circuitry ofFIG. 2.

Because the transimpedance amplifier shown FIG. 3 uses a pnp transistorinstead of a npn transistor for the transistor 204 (Q₂), the AC contentor component of the photodiode current will be absorbed or attenuated bythe transistor 204 when the impedance seen at the emitter of thetransistor 204 decreases. This is the case when the photodiode currentincreases and the optical signal detected by the photodiode hasincreased power.

Furthermore, the pnp transistor 204 can be replaced with an npntransistor as long as the input signal from the photodiode is notconnected at the collector of the npn transistor. The input signal isconnected with the emitter of the npn transistor. Also the cathode ofthe photodiode connector is connected with the emitter of the npntransistor in this embodiment.

The variation of the input impedance at the emitter of the transistor204 with the average photodiode current provides an automatic control ofthe transimpedance gain of the receiver with the average photodiodecurrent. In contrast, when an npn transistor is utilized instead of apnp transistor in the embodiment of FIG. 3 and the collector of the npntransistor is connected to the anode of the photodiode, the base o theoutput of 214 and the emitter to ground. The AC component of thephotodiode current is not attenuated because the impedance seen at thecollector of the npn transistor is not dependent on the average photodiode current. In addition, an npn transistor introduces increased openloop gain in the frequency response as the average photodiode currentincreases. The pnp transistor 204 does not introduce the gain that wouldotherwise be introduced by an npn transistor.

FIG. 4 illustrates another example of the variable impedance circuitry140. In this example, a npn transistor 302 is coupled with the pnptransistor 204. More specifically, the emitter of the transistor 302 isconnected to the emitter of the transistor 204. This permits thevariable impedance circuitry 140 to either source or sink the DC and ACcomponents of current and the photodiode can therefore be connected toeither a negative supply (or ground) or a positive supply. If thephotodiode 306 is connected to a negative supply or ground, then the npntransistor 302 has a variable impedance that depends on the averagecurrent of the photodiode 306. When the photodiode 304 is utilized, thenthe pnp transistor 204 has a variable impedance that is used forautomatic gain control through AC attenuation and DC cancellation.

With reference to FIGS. 2 and 3, the feedback circuit includes a verylow gain-bandwidth op-amp (B(s)) driving the base of the transistor 204and/or the transistor 302 (Q₂₂). The feedback circuit senses the DCoffset at the output of the A₁ gain stage (210) or at the output of thetransistors 206 and 208, or other suitable location. Because the gainstage of the amplifier 210 is DC coupled to the input stage of thetransimpedance amplifier, any offset between the transistor 200 andtransistor 202 collector voltages resulting from a difference ofcollector current is compensated by the transistor 302 sourcing currentat the input of the transimpedance amplifier or the transistor 204sinking current at the input of the transimpedance amplifier.

As a result, the feedback loop or circuit 130 from the amplifier or gainstage to the input of the transimpedance amplifier removes the DCcurrent or low frequency component of the photodiode signal. Therefore,the transconductance g_(m2) of the transistor 204 is proportional to theaverage photodiode current and hence the average received optical power(assuming the internal offset generated by the transimpedance amplifieris ignored).

For the transistor 302 (Q₂₂) or the transistor 204 (Q₂) thetransconductance is: $\begin{matrix}{{g_{m2} = \frac{I_{PD}}{V_{T}}},} & (1)\end{matrix}$

where I_(PD) is the average current of the photodiode or other opticaldevice that converts an optical signal in to an electrical signal suchas current.

In the frequency range where the transimpedance amplifier junctioncapacitances and the photodiode input capacitance can be ignored, theclosed loop transimpedance transfer function is given by:$\begin{matrix}{\frac{V_{out}}{I_{i\quad n}} = \frac{\frac{g_{m1}}{g_{m1} + g_{m2}}\quad {R_{C} \cdot A_{1}}}{1 + {\frac{g_{m1}}{g_{m1} + g_{m2}}\quad {R_{C} \cdot A_{1} \cdot g_{m2} \cdot B}\quad (s)}}} & (2)\end{matrix}$

or $\begin{matrix}{\frac{V_{out}}{I_{i\quad n}} = \frac{A}{1 + {A\quad \beta}}} & (3)\end{matrix}$

where the forward gain$A = {\frac{g_{m1}}{g_{m1} + g_{m2}}\quad {R_{C} \cdot A_{1}}}$

where and the feedback gain β=g_(m2)·B(s). The transconductance of thecommon base input stage is g_(m1) and is set by the base voltage and theresistor R_(E) 201 in the emitter.

In this example, a low frequency dominant pole OP-AMP with a DC gain ofB drives the base of the feedback transistors (pnp transistor 204 (Q₂)and/or the npn transistor 302 (Q₂₂)). The feedback gain can be written:$\begin{matrix}{\beta = {g_{m2}\quad \frac{B}{1 + \frac{s}{w_{0}}}}} & (4)\end{matrix}$

The transconductance of the transistors 200 and 202 depend on thevoltage V_(BASE) and the resistor R_(E) in series with their emitters.The bias of the input stage (I_(C(Q1)) and I_(C(Q0))) should beoptimized for bandwidth and noise. The bias of the input stage does notdepend on the average photodiode current and remains constant when theoptical power received at the photodiode changes.

The transimpedance amplifier 120 is examined below from the perspectivesof low optical power and of high optical power. At low optical power,g_(m2)<<g_(m1)(or I_(PD)<<I_(C(Q1))). Therefore, the transimpedance ofthe transimpedance amplifier transfer function can be simplified:$\begin{matrix}{\frac{V_{out}}{I_{i\quad n}} = \frac{R_{C} \cdot A_{1}}{1 + {{R_{C} \cdot A_{1} \cdot g_{m2} \cdot B}\quad (s)}}} & (5)\end{matrix}$

In the signal frequency band at low optical power, the transimpedancevalue of the transimpedance amplifier becomes: $\begin{matrix}{\frac{V_{out}}{I_{i\quad n}} = {R_{C} \cdot A_{1}}} & (6)\end{matrix}$

At high optical power, where g_(m1)<<g_(m2)(or I_(PD)>>I_(C(Q1)))$\begin{matrix}{\frac{V_{out}}{I_{i\quad n}} = \frac{\frac{g_{m1}}{g_{m2}}\quad {R_{C} \cdot A_{1}}}{1 + {{\frac{g_{m1}}{g_{m2}} \cdot R_{C} \cdot A_{1} \cdot g_{m2} \cdot B}\quad (s)}}} & (7)\end{matrix}$

In the signal frequency band at high optical power, the transimpedancevalue becomes: $\begin{matrix}{\frac{V_{out}}{I_{i\quad n}} = {\frac{g_{m1}}{g_{m2}} \cdot R_{C} \cdot A_{1}}} & (8)\end{matrix}$

or $\begin{matrix}{\frac{V_{out}}{I_{i\quad n}} = {\frac{I_{C\quad {({Q1})}}}{I_{PD}} \cdot R_{C} \cdot A_{1}}} & (9)\end{matrix}$

The low frequency feedback causes the closed loop gain frequencyresponse to have a low frequency cutoff given by: $\begin{matrix}{f_{{HPF} - {3d\quad B}} = {\frac{g_{m1}}{\left( {g_{m1} + g_{m2}} \right)} \cdot R_{C} \cdot A_{1} \cdot g_{m2} \cdot B \cdot f_{0}}} & (10)\end{matrix}$

where $f_{0} = \frac{w_{0}}{2\quad \pi}$

and B is the DC gain of the op-amp.

At low optical power, g_(m2)<<g_(m1)(or I_(PD)<<I_(C(Q1))). Thus,equation (10) can be simplified and the low frequency cutoff is givenby:

f _(HPF-3dB) =R _(C) ·A ₁ ·g _(m2) ·B·f ₀·  (11)

At high optical power, g_(m1)<<g_(m2)(or I_(PD)>>I_(C(Q1))) and equation(10) can be simplified and the low frequency cutoff is given by:

f _(HPF-3dB) =R _(C) ·A ₁ ·g _(m1) ·B·f ₀·  (12)

The low cutoff frequency at high optical power is not dependent on thetransistor 204 or on the transconductance of the transistor 204. The lowcutoff frequency is controlled. The low cutoff frequency represents the−3 dB low corner frequency in the frequency response of thetransimpedance amplifier. The present invention places a limit orcontrols the low corner frequency at high optical power.

In contrast, a similar analysis applied to a circuit that utilizes annpn transistor in place of the pnp transistor such that the collector ofthe npn transistor is connected with the input signal or current has alow frequency cutoff that is dependent on the transconductance of thenpn transistor. As the average photodiode current increases, the npntransistor causes the transimpedance amplifier to have a higher lowfrequency cutoff. One disadvantage is that a transimpedance amplifierusing an npn transistor in the place of the pnp transistor 204 is thatthe transimpedance amplifier does not function at lower frequencies forhigher optical power or larger input currents. The present invention,however, functions at lower frequencies for higher optical power orlarger input currents. This permits embodiments of the transimpedanceamplifier to be integrated with existing networks that operate at lowerfrequencies.

FIG. 5 is another embodiment of the automatic gain control low frequencyfeedback loop using a shunt feedback topology. The transistor 502 can bereplaced with the circuit illustrated in FIG. 4 to accommodate both anegative and positive supply as previously discussed.

The same analysis can be made for the shunt-feedback transimpedanceamplifier configurations that was made for the common base configurationof FIG. 3. Using nodal analysis on the small signal circuit lowfrequency model of the transimpedance amplifier input stageshunt-feedback amplifier in FIG. 5, the transimpedance transfer functionof the transimpedance amplifier can be extracted. In the frequency rangewhere the transimpedance amplifier junction capacitances and thephotodiode input capacitance can be ignored, the closed looptransimpedance transfer function is given by: $\begin{matrix}{\frac{V_{out}}{I_{i\quad n}} = {\frac{\frac{g_{m1} \cdot R_{C} \cdot A_{1} \cdot R_{F}}{1 + {g_{m1} \cdot R_{C} \cdot g_{m2} \cdot R_{F}}}}{1 + {{\frac{g_{m1} \cdot R_{C} \cdot R_{F}}{1 + {g_{m1} \cdot R_{C}} + {g_{m2} \cdot R_{F}}} \cdot g_{m2} \cdot A_{1} \cdot B}\quad (s)}}\quad {or}}} & (13) \\{\frac{V_{out}}{I_{i\quad n}} = {- \frac{A}{1 + {A \cdot \beta}}}} & (14)\end{matrix}$

where$A = \frac{g_{m1} \cdot R_{C} \cdot A_{1} \cdot R_{F}}{1 + {g_{m1} \cdot R_{C}} + {g_{m2} \cdot R_{F}}}$

(forward gain) and β=g_(m2)·B(s) (feedback gain).

At low optical power, where R_(C)·g_(m1)>>R_(F)·g_(m2) orR_(F)·I_(PD)<<R_(C)·I_(C(Q1)), the transimpedance of the transimpedanceamplifier transfer function can be simplified as: $\begin{matrix}{\frac{V_{out}}{I_{i\quad n}} = \frac{R_{F} \cdot A_{1}}{1 + {{R_{F} \cdot A_{1} \cdot g_{m2} \cdot B}\quad (s)}}} & (15)\end{matrix}$

In the signal frequency band, the transimpedance value becomes:$\begin{matrix}{\frac{V_{out}}{I_{i\quad n}} = {R_{F} \cdot A_{1}}} & (16)\end{matrix}$

At high optical power, where R_(C)·g_(m1)<<R_(F)·g_(m2) orR_(F)·I_(PD)>>R_(C)·I_(C(Q1)) $\begin{matrix}{\frac{V_{out}}{I_{i\quad n}} = \frac{\frac{g_{m1}}{g_{m2}} \cdot R_{C} \cdot A_{1}}{1 + {{\frac{g_{m1}}{g_{m2}} \cdot R_{C} \cdot A_{1} \cdot g_{m2} \cdot B}\quad (s)}}} & (17)\end{matrix}$

In the signal frequency band, the transimpedance value becomes:$\begin{matrix}{\frac{V_{out}}{I_{i\quad n}} = {\frac{g_{m1}}{g_{m2}} \cdot R_{C} \cdot A_{1}}} & (18)\end{matrix}$

or $\begin{matrix}{\frac{V_{out}}{I_{i\quad n}} = {\frac{I_{C\quad {({Q1})}}}{I_{PD}} \cdot R_{C} \cdot A_{1}}} & (19)\end{matrix}$

The low frequency feedback causes the closed loop gain frequencyresponse to have a low frequency cutoff given by: $\begin{matrix}{f_{{HPF} - {3d\quad B}} = {\frac{g_{m1} \cdot R_{C} \cdot A_{1} \cdot R_{F}}{1 + {g_{m1} \cdot R_{C}} + {g_{m2} \cdot R_{F}}} \cdot g_{m2} \cdot B \cdot f_{0}}} & (20)\end{matrix}$

where $f_{0} = \frac{w_{0}}{2\quad \pi}$

and B is the DC gain of the opamp.

At low optical power, R_(C)·g_(m1)>>R_(F)·g_(m2) orR_(F)·I_(PD)<<R_(C)·I_(C(Q1)). Therefore, equation (20) can besimplified and the low frequency cutoff is given by:

f _(HPF-3dB) =R _(F) ·A ₁ g _(m2) ·B·f ₀  (21)

At high optical power, R_(C)·g_(m1)<<R_(F)·g_(m2) orR_(F)·I_(PD)>>R_(C)·I_(C(Q1)). Therefore, equation (20) can besimplified and the low frequency cutoff is given by:

f _(HPF-3dB) =R _(C) ·A ₁ g _(m1) ·B·f ₀  (22)

Again, the low frequency cutoff at high optical power is not dependenton the transconductance of the transistor 502.

FIG. 6 illustrates an alternative embodiment of the variable impedancecircuitry using field effect transistors instead of bipolar junctiontransistors. The variable impedance circuitry can include many types offield effect transistors (MOSFETS, JFETS), BJT transistors, and the likeor any combination thereof. In another embodiment of the presentinvention, the photodiode that receives the optical signal is aphotodiode that also amplifies the optical signal or other device thatconverts the optical signal into an electrical signal or current.

FIG. 7 is a block diagram that plots the transimpedance of atransimpedance amplifier versus the average current of the photodiode.The plot 700 represents the transimpedance of an existing transimpedanceamplifier that utilizes, in one embodiment, an npn transistor for DCcancellation. For the plot 700, the collector of the npn transistor istypically connected with the input signal.

As the average photodiode current increases, the transimpedance ofexisting transimpedance amplifiers as illustrated by the plot 700 isrelatively steady and does not drop until the current gets larger(generally because of the saturation of the input stage). The plot 702,on the other hand, illustrates that the transimpedance of thetransimpedance amplifier illustrated in FIG. 3 decreases as the averagephotodiode current increases. The plot 702 also illustrates that thetransimpedance increases as the average photodiode current decreases. Atthe point 704, for example, the transimpedance illustrated in the plot700 is much higher than the transimpedance of the plot 702.

The variable impedance circuitry of the present invention, whichincludes a pnp transistor in one embodiment, enables the transimpedanceamplifier to adjust to the transimpedance gain gradually as the averagephotodiode current increases or decreases.

FIG. 8 is a graph that plots the low corner frequency of thetransimpedance amplifier as it varies with the average photodiodecurrent. The plot 800 illustrates a plot of the low corner frequency ofexisting transimpedance amplifiers. The plot 802 illustrates a plot ofthe low corner frequency of a transimpedance amplifier in accordancewith the present invention. More particularly, the plot 802 representsthe low corner frequency of the transimpedance amplifier illustrated inFIG. 3.

FIG. 8 illustrates that as the average photodiode current increases, thelow cutoff frequency of the plot 800 increases rapidly and linearly. Aspreviously described, this can ultimately result in circuit failure. Inother words, the capability to transmit lower data rates at high inputpower optical sensitivity of the transimpedance amplifier represented bythe plot 800 is diminished. The diminished transmission capabilityresults because the low cutoff frequency is not controlled and increasesquickly as the input power of the optical signal or of the current fromthe optical device that converts the optical signal to an input currentincreases.

In contrast, the low cutoff frequency of the transimpedance amplifierrepresented by the plot 802 levels off or approaches an upper limit asthe average photodiode current increases. Because the increase in thelow cutoff frequency in the plot 802 is substantially less than theincrease illustrated by the plot 800, the transimpedance amplifierrepresented by the plot 802 can successfully accommodate a larger rangeof data rates. The optical sensitivity is improved and thetransimpedance amplifier can interact with legacy systems that mayoperate at lower frequencies.

FIG. 9 illustrates another embodiment of a transimpedance amplifier withautomatic gain control and DC cancellation circuitry. FIG. 9 is similarto FIG. 5 and also includes transistor 907. Transistors 902 and 907 arepnp transistors. The shunt-feedback transimpedance amplifier of FIG. 9can be used to amplify signals having a wide range of different opticalpowers. Using nodal analysis on the circuit of FIG. 9, thetransimpedance transfer function can be identified. In FIG. 9,transistor 907 can be some factor N smaller in the emitter size thantransistor 902. Accordingly, since amplifier 906 cancels DC signals:

I ₂ =I _(PD) +I _(RF1)  (23)

and $\begin{matrix}{I_{2} = {I_{PD} + {I_{RF1}\quad {and}}}} & (23) \\{I_{3} = {\frac{I_{2}}{N}\quad {furthermore}}} & (24) \\{I_{RF5} = {I_{3}\quad {and}}} & (25) \\{{R_{F5} \cdot I_{RF5}} = {{R_{F1} \cdot I_{RF1}}\quad {and}\quad {thus}}} & (26) \\{I_{RF5} = {I_{RF1} = I_{3}}} & (27)\end{matrix}$

furthermore

I_(RF5)=I₃  (25)

and

 R _(F5) ·I _(RF5) =R _(F1) ·I _(RF1)  (26)

and thus

I_(RF5)=I_(RF1)=I₃  (27)

From equations (23)-(27) it can be calculated that: $\begin{matrix}{I_{RF5} = {I_{RF1} = \frac{I_{PD}}{\left( {N - 1} \right)}}} & (28)\end{matrix}$

Additionally, since amplifier 906 cancels DC signals: $\begin{matrix}{V_{A} = {V_{B} = {V_{BEQ1} + {R_{F} \cdot \frac{I_{PD}}{\left( {N - 1} \right)}} + V_{BEQ5}}}} & (29)\end{matrix}$

and thus I_(Q1) (the current through Q₁) is given by: $\begin{matrix}{I_{Q1} = {\frac{V_{CC} - V_{A}}{R_{C}} = {\frac{V_{CC} - {2V_{BE}} - {R_{F} \cdot \frac{I_{PD}}{\left( {N - 1} \right)}}}{R_{C}} = {\frac{V_{CC} - {2V_{BE}}}{R_{C}} - {\left( \frac{R_{F}}{R_{C}} \right) \cdot \left( \frac{I_{PD}}{\left( {N - 1} \right)} \right)}}}}} & (30)\end{matrix}$

Thus, I_(Q1) decreases when the photodiode average current increases

The transconductance g_(m1) of transistor 901 (Q₁) is given by:$\begin{matrix}{g_{m1} = {\frac{I_{Q1}}{V_{T}} = {\left\lbrack {\frac{V_{CC} - {2V_{BE}}}{R_{C}} \cdot \left( \frac{1}{R_{C}} \right)} \right\rbrack - \left\lbrack {\left( \frac{R_{F}}{R_{C}} \right) \cdot \left( \frac{I_{PD}}{\left( {N - 1} \right)} \right) \cdot \left( \frac{1}{V_{T}} \right)} \right\rbrack}}} & (31)\end{matrix}$

At high optical power (and similar to equation 22), the low frequencycutoff is given by:

f _(HPF-3dB) =R _(C) ·g _(m1) ·B·f ₀  (32)

Similar to FIG. 5, the low frequency cutoff at high optical power is notdependent on the transconductance of the transistor 902. Further, thelow frequency cutoff can be adjusted by varying the value of N.Accordingly, the transconductance amplifier of FIG. 9 can providesufficient gain at a variety of frequencies.

FIG. 10 is an example plot of the gain of a transimpedance amplifierversus frequency. As depicted in FIG. 10, the circuit of FIG. 9 providessimilar gain over a range of frequencies from frequency 1001 tofrequency 1002. In one embodiment, frequency 1001 could be set toaccommodate a data rate of 50 Mb/s and frequency 1002 could be set toaccommodate a data rate of 2.5 Gb/s (or an even higher data rate). Thus,similar gain is provided at data rates ranging from (and including) 50MB/s to 2.5 GB/s (or an even higher data rate).

FIG. 11 is an example of a number of plots of transimpedance amplifiertransimpedance gain versus photodiode average current. Plot 1101 depictsan example of transimpedance amplifier transimpedance gain versusphotodiode average current for a prior art circuit. Plot 1102 depicts anexample of transimpedance amplifier transimpedance gain versusphotodiode average current for a circuit (e.g., the circuit of FIG. 5)that uses a pnp transistor (e.g., transistor 502) for gain control. Plot1103 depicts an example of transimpedance amplifier transimpedance gainversus photodiode average current for a circuit that uses two pnptransistors, having an emitter size ratio M of 10, for gain control. Forexample, plot 1103 could represent the circuit of FIG. 9 where theemitter size of 902 divided by the emitter size of 907 equals 10.

Plot 1104 depicts an example of transimpedance amplifier transimpedancegain versus photodiode average current for a circuit that uses two pnptransistors, having an emitter size ratio M of 5, for gain control. Forexample, plot 1104 could represent the circuit of FIG. 9 where theemitter size of 902 divided by the emitter size of 907 equals 5. Plot1105 depicts an example of transimpedance amplifier transimpedance gainversus photodiode average current for a circuit that uses two pnptransistors, having an emitter size ratio M of 2.5, for gain control.For example, plot 1105 could represent the circuit of FIG. 9 where theemitter size of 902 divided by the emitter size of 907 equals 2.5.

As depicted in FIG. 11, as photodiode average current increases, thetransimpedance gain for circuits that utilize two pnp transistors forgain control (e.g., the circuit of FIG. 9) decreases more quickly thanthe transimpedance gain for prior art circuits and for circuits thatutilize one pnp transistor for gain control (e.g., the circuit of FIG.5). For example, plots 1103, 1104, and 1105 decrease more quickly thanplot 1102. Further, as photodiode average current increases, thetransimpedance gain for circuits having lower emitter size ratiosdecreases more quickly than the transimpedance gain for circuits havinggreater emitter size ratios. For example, plot 1105 (M=2.5) decreasesmore quickly than both plot 1103 (M=10) and plot 1104 (M=5).

FIG. 12 is an example of a number of plots of transimpedance amplifierlow frequency cut-off versus photodiode average current. Plot 1201depicts an example of transimpedance amplifier low frequency cut-offversus photodiode average current for a prior art circuit. Plot 1202depicts an example of transimpedance amplifier low frequency cut-offversus photodiode average current for a circuit (e.g., the circuit ofFIG. 5) that uses a pnp transistor (e.g., transistor 502) for gaincontrol. Plot 1203 depicts an example of transimpedance amplifier lowfrequency cut-off versus photodiode average current for a circuit thatuses two pnp transistors, having an emitter size ratio M of 10, for gaincontrol. For example, plot 1203 could represent the circuit of FIG. 9where the emitter size of 902 divided by the emitter size of 907 equals10.

Plot 1204 depicts an example of transimpedance amplifier low frequencycut-off versus photodiode average current for a circuit that uses twopnp transistors, having an emitter size ratio M of 5, for gain control.For example, plot 1204 could represent the circuit of FIG. 9 where theemitter size of 902 divided by the emitter size of 907 equals 5. Plot1205 depicts an example of transimpedance amplifier low frequencycut-off versus photodiode average current for a circuit that uses twopnp transistors, having an emitter size ratio M of 2.5, for gaincontrol. For example, plot 1203 could represent the circuit of FIG. 9where the emitter size of 902 divided by the emitter size of 907 equals2.5.

As depicted in FIG. 12, as photodiode average current increases, the lowfrequency cut-off for circuits that utilize two pnp transistors for gaincontrol (e.g., the circuit of FIG. 9) is more controlled than the lowfrequency cut-off for prior art circuits and for circuits that utilizeone pnp transistor for gain control (e.g., the circuit of FIG. 5). Forexample, plots 1203, 1204, and 1205 begin transition lower oncephotodiode average current reaches a threshold value. Further, asphotodiode average current increases, the low frequency cut-off forcircuits having lower emitter size ratios transitions to decreasing morequickly than for circuits having greater emitter size ratios. Forexample, plot 1205 (M=2.5) transitions to decreasing more quickly thanboth plot 1203 (M=10) and plot 1204 (M=5).

The present invention may be embodied in other specific forms withoutdeparting from its spirit or essential characteristics. The describedembodiments are to be considered in all respects only as illustrativeand not restrictive. The scope of the invention is, therefore, indicatedby the appended claims rather than by the foregoing description. Allchanges which come within the meaning and range of equivalency of theclaims are to be embraced within their scope.

What is claimed is:
 1. A transimpedance amplifier circuit with anamplifier input, the transimpedance amplifier circuit having acontrolled low cutoff frequency as average input current to theamplifier input increases, the transimpedance amplifier circuitcomprising: an optical device having an optical device input terminaland an optical device output terminal, the optical device for receivingan optical signal at the optical device input terminal, converting theoptical signal to an input current, and providing the input current atthe optical device output terminal; a forward transimpedance circuitconnected to the optical device output terminal, the forwardtransimpedance circuit for receiving the input current and generating anoutput signal based on the input current; a feedback circuit thatincludes: a first circuit that for detecting a low frequency componentof the output signal; and a second circuit that is driven by the lowfrequency component of the output signal and is connected to the forwardtransimpedance circuit such that the impedance of the second circuitpresented at the amplifier input decreases as the output signalincreases, the second circuit including: a first pnp transistor having afirst base terminal and a first emitter terminal, wherein the firstemitter terminal is connected to the amplifier input; and a second pnptransistor having a second base terminal, the second base terminal beingconnected to the first base terminal, the second pnp transistor havingan emitter size that is some factor smaller than an emitter size of thefirst pnp transistor.
 2. A transimpedance amplifier circuit as recitedin claim 1, wherein an impedance seen at the first emitter terminal isdependent on the average current of the input current and wherein thelow cutoff frequency does not increase linearly as the input currentincreases.
 3. A transimpedance amplifier circuit as recited in claim 1,wherein the second circuit has variable impedance such that increasingan optical overload of the transimpedance amplifier does not diminish anoptical sensitivity of the transimpedance amplifier.
 4. A transimpedanceamplifier circuit as recited in claim 1, wherein the first circuit andthe second circuit shunt a DC component of the input current such that aDC component of the output signal is significantly reduced.
 5. Atransimpedance amplifier circuit as recited in claim 1, wherein thefirst circuit includes a low frequency operational amplifier.
 6. Atransimpedance amplifier with an amplifier input, the transimpedanceamplifier having a controlled low cutoff frequency as average input tothe transimpedance amplifier increases, the transimpedance amplifiercomprising: an input stage for receiving an input current signalprovided at the output terminal of an optical device, the input stagegenerating an output voltage based on the received input current; a gainstage for amplifying the output voltage to generate an amplified signal;and a feedback circuit that includes: a low frequency circuit fordetecting a low frequency component of the amplified signal such thatthe low frequency component can be removed from the amplified signal;and variable impedance circuitry, wherein an impedance of the variableimpedance circuitry is dependent on an average current of the inputcurrent signal such that the impedance decreases as the average currentincreases and wherein a low cutoff frequency of the transimpedanceamplifier decreases when the average current increases to greater than aspecified threshold, the variable impedance circuitry including: a firstpnp transistor having a first base terminal and a first emitterterminal, wherein the emitter terminal is connected to the amplifierinput, the first pnp transistor have a first emitter size; and a secondpnp transistor having a second base terminal, the second base terminalbeing connected to the first base terminal, the second pnp transistorhaving a second emitter size that is some factor smaller than the firstemitter size.
 7. A transimpedance amplifier as recited in claim 6,wherein the input stage is in a shunt feedback configuration and whereinthe gain stage is an amplifier.
 8. A transimpedance amplifier as recitedin claim 6, wherein the low frequency circuit further comprises a lowfrequency operational amplifier.
 9. A transimpedance amplifier asrecited in claim 6, wherein the low frequency circuit detects andreduces the low frequency component at the input stage by shunting thelow frequency component of the input current signal.
 10. Atransimpedance amplifier as recited in claim 6, wherein the first pnptransistor that has a transconductance that does not affect the lowcutoff frequency of the transimpedance amplifier as the input currentsignal increases.
 11. In a system that receives input currents ofdifferent magnitudes, the system including a forward transimpedancecircuit, a low frequency detection circuit, a variable impedancecircuit, a first pnp transistor, and a second pnp transistor, the firstpnp transistor having a first base terminal and a first emitter size,the second pnp transistor having a second base terminal, the second baseterminal being connected to the first base terminal, the second pnptransistor having a second emitter size that is some factor smaller thanthe first emitter size, a method for controlling a low cutoff frequencyas average input current to the system increases, the method comprising:the forward transimpedance circuit receiving an input current from anoptical device; the forward transimpedance circuit generating an outputsignal based on the input current; the low frequency detection circuitdetecting a low frequency component of the output signal; utilizing thelow frequency component to determine the impedance of the variableimpedance circuit such that the impedance of the of the variableimpedance circuit decreases as the average input current increases;utilizing the first and second pnp transistors to control the low cutofffrequency such that the low cutoff frequency transitions to decreasingwhen the magnitude of average input current reaches a specifiedthreshold.